DC—DC converters providing reduced deadtime

ABSTRACT

A DC-DC power converter ( 100 ) that provides increased power density, reduced size, and reduced costs of manufacture. The DC-DC power converter ( 100 ) includes first and second input terminals ( 101, 102 ), a plurality of output terminals ( 116, 117 ), and at least one electrical element ( 108/109 ) connected to at least one of the first and second input terminals ( 101, 102 ). In the event a first voltage is applied across the first and second input terminals ( 101, 102 ), the electrical element ( 108/109 ) provides a second voltage having a value between the first voltage value and a reference voltage value. The DC-DC power converter ( 100 ) further includes a transformer ( 110 ) having a primary winding ( 111 ) and a secondary winding ( 112 ), and a switch assembly ( 103–107 ) operatively connected to the first input terminal ( 101 ), the second input terminal ( 102 ), the electrical element ( 108/109 ), and the transformer primary winding ( 111 ). In the event the first voltage is applied across the first and second input terminals ( 101, 102 ), the switch assembly ( 103–107 ) switchably applies the first, second, and reference voltages across the transformer primary winding ( 111 ) to generate at least one third voltage across the transformer secondary winding ( 112 ). The DC-DC power converter ( 100 ) further includes a rectifier ( 113 ) connected between the transformer secondary winding ( 112 ) and the output terminals ( 116, 117 ).

CROSS REFERENCE TO RELATED APPLICATIONS

This application claims priority of U.S. Provisional Patent ApplicationNo. 60/343,607 filed Dec. 28, 2001 entitled DUAL BRIDGE CONVERTER, andU.S. Provisional Patent Application No. 60/430,585 filed Dec. 3, 2002entitled THREE-LEVEL FAST TRANSIENT LOW OUTPUT VOLTAGE DC-DC CONVERTERS.

STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT

N/A

BACKGROUND OF THE INVENTION

The present invention relates generally to power converters such asDC-DC power converters, and more specifically to DC-DC power convertersconfigured to provide reduced deadtime.

DC-DC power converters are known for converting an input DC voltage intoan output DC voltage having a value smaller or larger than the inputvoltage value, possibly with opposite polarity and/or with isolation ofthe input and output ground references. DC-DC converters normally acceptinput energy from a voltage source at a voltage input, and provideconverted output energy at a voltage (and current) output, which isusually a filtered output that operates as a voltage sink.

When isolation is employed in a DC-DC converter, the input voltage istypically switched on and off at a high frequency, and provided to atransformer, which provides the input/output isolation and the suitablevoltage conversion. However, because the input voltage is switched atthe high frequency, the output voltage and current typically cannot bedirectly provided to a load in a regulated manner. An inductor isgenerally required in the energy conversion to act as a current filter.The size and value of the inductor are often critical to meeting theperformance specifications. A large inductance volume normally reducesthe power density of the converter. Further, because inductors withlarge inductance values have low slew rates, the response time of theconverter to load current disturbances is slowed down. Accordingly,smaller inductance volumes and values are desirable.

Isolated DC-DC converters typically operate with at least some amount ofdeadtime. For example, a conventional full-bridge converter has deadtimeduring its operation. Besides preventing switches in the same leg of theconverter from conducting simultaneously, this deadtime allowsconventional dual-end (e.g., half-bridge, full-bridge, push-pull, etc.)converters to have a regulated output voltage when the input voltagechanges.

During the deadtime, the energy into the input is discontinuous, causinga large input current ripple. Large input filters are therefore employedto satisfy conducted Electromagnetic Compatibility (EMC) requirements.This deadtime also necessitates a large output inductor to smooth theoutput voltage, and to limit the current ripple through it. However, thelarge output inductor slows the output response time. The volume of theoutput inductor also takes up valuable board space. Further, as thelength of the deadtime increases, the size of the output inductor oftenincreases. Because of this deadtime, simple self-driven synchronousrectification schemes typically cannot be used to achieve highefficiency of power conversion in low voltage, high current output DC-DCconverter applications.

Certain topologies produce little or no deadtime, which means thatenergy is continuously transmitted from the input DC source to theoutput load during the entire switching period. Other topologies mayprovide a reduced deadtime. Because the input and output current ripplesare generally lower in DC-DC converters having reduced or no deadtime,the input filter is generally smaller. Further, the lower outputinductance value improves the output transient speed and reduces theoutput filter size, thereby improving the power density and outputtransient response of DC-DC converter. Moreover, the peak to peakvoltage ripple across the inductor generally decreases, which allows areduced inductor volume. Conventional techniques for reducing deadtimeinclude magnetic transformer tapping, and two transformerimplementations. However, magnetic transformer tapping typically hasmanufacturability problems, which can lead to difficulties intransformer operation such as high leakage inductance or magnetic fluximbalance. In addition, the extra switches employed in magnetictransformer tapping can increase losses. Further, the two transformerimplementation typically requires an additional magnetic core, whichtakes up valuable board space. The power density of such conventionalDC-DC converter implementations may also be reduced.

Accordingly, there is a continuing need to develop and improve DC-DCconverters that operate with reduced or no deadtime.

BRIEF SUMMARY OF THE INVENTION

In accordance with the present invention, DC-DC power converters aredisclosed that provide increased power density, reduced size, andreduced costs of manufacture. Benefits of the presently disclosed DC-DCpower converters are achieved at least in part by reducing oreliminating the amount of deadtime during power converter operation.

In one embodiment, a DC-DC power converter includes first and secondinput terminals, a plurality of output terminals, and at least oneelectrical element connected to at least one of the first and secondinput terminals. The electrical element is operative, in the event afirst voltage is applied across the first and second input terminals, toprovide a second voltage having a value between the first voltage valueand a reference voltage value. The DC-DC power converter furtherincludes a transformer having at least one primary winding and at leastone secondary winding, and a switch assembly having a plurality ofswitching elements operatively connected to the first input terminal,the second input terminal, the electrical element, and the transformerprimary winding. The switch assembly is operative, in the event thefirst voltage is applied across the first and second input terminals, toswitchably apply the first voltage, the second voltage, and thereference voltage, across the transformer primary winding to generate atleast one third voltage across the transformer secondary winding. TheDC-DC power converter further includes a rectifier connected between thetransformer secondary winding and the output terminals.

In a second embodiment, a DC-DC power converter includes a switchassembly, and a second DC-DC power converter operatively connected tothe switch assembly. The switch assembly includes first and second inputterminals, first and second output terminals, and at least oneelectrical element connected across the first and second inputterminals. The electrical element is operative, in the event a firstvoltage is applied across the first and second input terminals, toprovide a second voltage having a value between the first voltage valueand a reference voltage value. The switch assembly further includes aswitch subassembly having a plurality of switching elements. The switchsubassembly is operatively connected to the first input terminal, thesecond input terminal, and the electrical element. Further, the switchsubassembly is operative, in the event the first voltage is appliedacross the first and second input terminals, to switchably apply thefirst voltage, the second voltage, and the reference voltage across thefirst and second output terminals. The second DC-DC power converter isoperatively connected to the first and second output terminals of theswitch assembly, and configured to receive the first voltage, the secondvoltage, and the reference voltage applied across the first and secondoutput terminals of the switch assembly, and to generate at least onethird voltage.

By applying the above-described first voltage having a maximum valueless than or equal to two times the first voltage minimum value, whileoperating the respective switching elements according to at least onepredetermined duty ratio, the DC-DC power converter can be made tooperate with no deadtime. Further, by applying the first voltage havinga maximum value greater than two times the first voltage minimum value,while operating the respective switching elements according to at leastone predetermined ratio, the DC-DC power converter can be made tooperate with a predetermined amount of deadtime.

Other features, functions, and aspects of the invention will be evidentfrom the Detailed Description of the Invention that follows.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWING

The invention will be more fully understood with reference to thefollowing Detailed Description of the Invention in conjunction with thedrawings of which:

FIG. 1 is a schematic diagram of a full-bridge DC-DC converter accordingto the present invention;

FIG. 2 is a timing diagram illustrating circuit waveforms produced bythe converter of FIG. 1;

FIG. 3 a is a schematic diagram of a first alternative embodiment of theconverter of FIG. 1;

FIG. 3 b is a timing diagram illustrating circuit waveforms produced bythe converter of FIG. 3 a;

FIG. 4 is a schematic diagram of a second alternative embodiment of theconverter of FIG. 1;

FIG. 5 is a timing diagram illustrating circuit waveforms produced bythe converter of FIG. 4;

FIG. 6 is a schematic diagram of a third alternative embodiment of theconverter of FIG. 1;

FIG. 7 is a timing diagram illustrating circuit waveforms produced bythe converter of FIG. 6;

FIG. 8 is a schematic diagram of a fourth alternative embodiment of theconverter of FIG. 1;

FIGS. 9 a–9 b are schematic diagrams of electrical components andconnections employed in the converters of FIGS. 1, 4, 6, and 8;

FIG. 10 is a schematic diagram of a push-pull DC-DC converter coupled toa three-level switch cell according to the present invention;

FIG. 11 is a timing diagram illustrating circuit waveforms produced bythe converter of FIG. 10;

FIG. 12 is a schematic diagram of the three-level switch cell of FIG. 10coupled to a generalized DC-DC converter;

FIG. 13 a is a schematic diagram of the three-level switch cell of FIG.10 coupled to a full-bridge DC-DC converter;

FIG. 13 b is a schematic diagram of a dual-forward converter employingthe three-level switch cell of FIG. 10;

FIG. 13 c is a schematic diagram of a half-bridge converter employingthe three-level switch cell of FIG. 10;

FIG. 14 is a schematic diagram of an isolated single-ended forward DC-DCconverter employing the three-level switch cell of FIG. 10;

FIG. 15 is a timing diagram illustrating circuit waveforms produced bythe converter of FIG. 14;

FIG. 16 a is a schematic diagram of a current-fed push-pull buck DC-DCconverter;

FIG. 16 b is a schematic diagram of a current fed full-bridge DC-DCconverter;

FIG. 16 c is a schematic diagram of a current fed dual-forward DC-DCconverter;

FIG. 17 is a schematic diagram of a current fed half-bridge DC-DCconverter employing the three level switch cell of FIG. 10;

FIG. 18 is a schematic diagram of a three-level buck DC-DC converteremploying the three-level switch cell of FIG. 10;

FIG. 19 is a block diagram of the three-level buck DC-DC converter ofFIG. 18 coupled to another DC-DC converter;

FIG. 20 is a detailed view of the DC-DC converter coupled to thethree-level buck DC-DC converter of FIG. 18;

FIG. 21 a is a schematic diagram of a three-level two-stage buck andpush-pull DC-DC converter;

FIG. 21 b is a schematic diagram of an alternative embodiment of athree-level two-stage buck and push-pull DC-DC converter;

FIG. 21 c is a schematic diagram of a three-level two-stage full-bridgeDC-DC converter;

FIG. 21 d is a schematic diagram of an alternative embodiment of athree-level two-stage full-bridge DC-DC converter;

FIG. 21 e is a schematic diagram of a three-level two-stage dual-forwardconverter;

FIG. 21 f is a schematic diagram of an alternative embodiment of athree-level two-stage dual-forward converter;

FIG. 21 g is a schematic diagram of a three-level two-stage half-bridgeconverter;

FIG. 22 a is a schematic diagram of a four-level switch cell;

FIG. 22 b is a schematic diagram of the switch cell of FIG. 22 ageneralized for (n_(odd)+1) levels; and

FIG. 22 c is a schematic diagram of the switch cell of FIG. 22 ageneralized for (n_(even)+1) levels.

DETAILED DESCRIPTION OF THE INVENTION

U.S. Provisional Patent Application No. 60/343,607 filed Dec. 28, 2001entitled DUAL BRIDGE CONVERTER, and U.S. Provisional Patent ApplicationNo. 60/430,585 filed Dec. 3, 2002 entitled THREE-LEVEL FAST TRANSIENTLOW OUTPUT VOLTAGE DC-DC CONVERTERS, are incorporated herein byreference.

DC-DC power converters are disclosed that provide increased powerdensity, reduced size, and reduced costs of manufacture. The presentlydisclosed DC-DC power converters are configured to substantially reduceor eliminate the amount of deadtime occurring during power converteroperation.

FIG. 1 depicts an illustrative embodiment of a DC-DC power converter100, in accordance with the present invention. In the illustratedembodiment, the converter 100 comprises a full-bridge converterincluding a plurality of switching elements 103–107. The switchingelements 103–104 are connected in series across input terminals 101–102,and form a node 120. The switching elements 105–106 are also connectedin series across the input terminals 101 and 102, and form a node 119.The converter 100 further comprises a transformer 110 having a primarywinding 111 and a secondary winding 112, and electrical elements108–109. The primary transformer winding 111 is connected across thenodes 119–120. The electrical elements 108–109 are connected in seriesacross the input terminals 101–102, and form a node 118. The switchingelement 107 connects the nodes 118–119. In the presently disclosedembodiment, the electrical elements 108–109 comprise respectivecapacitors. It is understood, however, that the electrical elements108–109 may alternatively comprise respective batteries, DC-DCconverters, or any other suitable electrical and/or electroniccomponents or devices.

It is noted that the phrase “switching element” is employed herein torepresent one or more electronic components (e.g., switching transistorsand diodes) that (1) when turned “on”, allow current to pass in one ortwo directions, and (2) when turned “off”, block current in at least onedirection. Any suitable combination of switching elements such astransistors or diodes may be employed to achieve the desired switchingoperation. For example, a diode is a switching element that carriescurrent in one direction, and blocks current in the opposite direction.A Field Effect Transistor (FET) carries current in both directions whenturned on, but blocks current in only one direction when turned off. TwoFETs connected in series, with the source of the first FET connected tothe source of the second FET, can carry or block current in eitherdirection. A diode in series with a FET can carry current in onedirection, and block current bi-directionally. Other suitable switchingelements (e.g., Silicon Controlled Rectifiers (SCRs), TRIACS,thyristors, etc.) and combinations thereof may also be employed. Thediode is a two terminal device that does not use a separate controlsignal to determine its switching action. Other switching elements suchas FETs are three terminal devices that employ a control signal todetermine the timing for turning the element on and off. The choice ofwhich switching elements or combinations thereof to use is generallydetermined by the conducting, blocking, turn-on, and turn-offrequirements of the switching element within the target circuitapplication.

As shown in FIG. 1, the DC-DC converter 100 further includes a rectifier113 and a filtering circuit 114, which in the illustrated embodimentcomprises a low pass filter. The voltage across the secondary winding112 is applied to the rectifier 113 to obtain a rectified voltageV_(rect), which is then applied to the filtering circuit 114. The outputvoltage of the converter 100 is the filtered output voltage V_(o) takenacross nodes 116–117. It is noted that all of the components of theconverter 100 of FIG. 1, including the switching elements 103–107, areconsidered to be ideal components for clarity of discussion.

FIG. 1 depicts one center tapped secondary winding 112 of thetransformer 110 connected to the two-diode full wave rectificationcircuit 113. This is a preferred approach among several possibleapproaches that may be used to obtain the rectified voltage V_(rect). Inalternative embodiments, other approaches may be employed including, butnot limited to, a secondary winding that is not center tapped withfour-diode bridge rectification, or a secondary winding that is notcenter tapped with one-diode half wave rectification. Accordingly, theconverter 100 may employ any suitable technique for connecting thetransformer secondary winding 112 (e.g., center tapped or not centertapped) to the rectification circuit 113 (e.g., half wave, full wave,bridge, or any other suitable rectification type).

Specifically, when the switching element 107 is open, the converter 100operates as a full bridge converter. When the switching element 107 isclosed, and the switching element 106 is open, the voltage at the node118 is approximately equal to the voltage across the capacitor 109. Inthis configuration, it is possible to create a voltage across theprimary winding 111 having a value between V_(in) and 0 volts. In theillustrated embodiment, the voltage across the capacitor 108 is equal tothe voltage across the capacitor 109. The voltage at the node 119 istherefore approximately equal to one half of the input voltage (i.e.,V_(in)/2). It is noted that in a conventional full bridge converter, thevoltage across the primary transformer winding switches between V_(in)and 0 volts. It is possible to operate the converter 100 when thesevoltages are not equal, provided that the operation maintains an ACsignal on the transformer 110 with no sustainable DC component.

FIG. 2 depicts illustrative circuit waveforms for the DC-DC converter100 (see FIG. 1). In the illustrated embodiment, a signal V₁ controlsthe switching element 103. Specifically, when V₁ is a high voltagelevel, the switching element 103 is closed, and current passes throughthe switch 103. When V₁ is a low voltage level, the switching element103 turns off and blocks the current. In a similar manner, a signal V₂controls the switching element 104, a signal V₃ controls the switchingelement 105, a signal V₄ controls the switching element 106, and asignal V₅ controls the switching element 107.

In the presently disclosed embodiment, the signals V₁ and V₂ depicted inFIG. 2 are two 50% duty ratio complementary control signals with aswitching frequency f. The signals V₃ and V₄ are control signals with aduty ratio of D, and a switching frequency f. The signal V₅ drives theswitching element 107 at an operating frequency of f₀=2f. It is notedthat the switching element 107 is turned on when both of the switchingelements 105–106 are turned off.

For example, suppose that the DC-DC converter 100 operates in steadystate, and its output inductor current is in the continuous conductionmode. As shown in FIG. 2, from time to to t₁, V₁ and V₄ are both highvoltage levels, and therefore the switching elements 103 and 106 areboth turned on. Further, the voltage V₁₁₁ of the transformer primarywinding 111 is approximately equal to the input voltage V_(in). Duringthe time t₀ to t₁, the input current i_(in), which is approximatelyequal to the primary winding current i₁₁₁, increases until it reachesi_(111,max) at time t₁.

From time t₁ to t₂, V₄ is a low voltage level and V₅ is a high voltagelevel, and therefore the switching element 106 is off and switchingelement 107 is on. Further, the primary winding voltage V₁₁₁ isapproximately equal to V_(in)/2, and the input current i_(in) decreasesfrom i_(111,max)/2. In addition, the current i₁₁₁, which decreases fromi_(111,max), is supplied by i_(in), the discharging current of thecapacitor 108, and the charging current of the capacitor 109.

From time t₂ to t₃, because V₁ and V₅ are low voltage levels, theswitching elements 103 and 107 both turn off. In addition, because V₂and V₃ are high voltage levels, the switching elements 104–105 turn on.After a short transient time, the primary winding voltage V₁₁₁ isapproximately equal to the negative input voltage, −V_(in). Further, theprimary winding current i₁₁₁ transitions to a negative current value,specifically, from a current value of i₁₁₁=i_(111,min) immediatelybefore time t₂, to a current value of I₁₁₁=−i_(111,min) just after timet₂. From time t₂ to t₃, i₁₁₁ changes from −i_(111,min) to −i_(111,max).

At time t₃, because V₃ is a low voltage level and V₅ is a high voltagelevel, the switching element 105 turns off and the switching element 107turns on. Further, the primary winding voltage V₁₁₁ is approximatelyequal to −V_(in)/2, and the input current i_(in) is approximately equalto i_(111,max)/2. After time t₃, the primary winding current changesfrom −i_(111,max) towards −i_(111,min), reaching −i_(111,min) at timet₄. From time t₃ to t₄, the primary winding current i₁₁₁ is supplied bythe input current i_(in), the charging current of the capacitor 108, andthe discharging current of the capacitor 109. From time t₄, after a veryshort transient time, V₁₁₁ and i₁₁₁ change polarity, and the processrepeats hereafter as described above.

The voltage on the secondary winding 112, i.e., V₁₁₂, is linearlyrelated to the voltage on the primary winding 111, i.e., V₁₁₁, asfollows:V₁₁₂=nV₁₁₁,  (1)in which “n” is the turns ratio n=n₁₁₂/n₁₁₁, n₁₁₂ is the number ofsecondary windings, and n₁₁₁ is the number or primary windings. Thesecondary winding voltage V₁₁₂ is rectified, and depicted as V_(rect) inFIG. 2. Assuming ideal rectification, V_(rect) is approximately equal tothe absolute value of the primary winding voltage V₁₁₁ times the turnsratio n. Further, the average value of V_(rect) is approximately equalto the DC output voltage V_(o) after it is sent through the filter 114.

When the DC-DC converter 100 operates in the above-described manner, theconverter 100 is in a “no deadtime” operational mode. That is, at anygiven time, energy is being transmitted from the input source to theoutput load (note that the switching transient time is negligiblecompared to the cycle time of operation). In this case, V_(rect)switches between nV_(in) and (n/2)V_(in). The output voltage isregulated by controlling the length of time V_(rect) is at the valuenV_(in), versus the length of time V_(rect) is at the value (n/2)V_(in).If the switching period of the Pulse Width Modulated (PWM) drivingsignals for the switching elements 103–107 is equal to 2T₀, then theduty ratio is equal to D₀=(t3−t2)/T0, which is indicative of theconduction time of the switching element 105. The duty ratio has a valuebetween 0 and 1 (equivalently referred to as being between 0% and 100%).The complementary duty ratio, 1−D₀, is indicative of the conduction timeof the switching element 107. Changing the value of D₀ changes theaverage value of V_(rect), so thatV ₀ =nV _(in)(1+D ₀)/2.  (2)

From equation (2), it can be seen that to maintain the property of nodeadtime operation, the input voltage V_(in) is limited within a 2:1voltage range. If the input voltage range is less than 2:1, then thetransformer turns ratio n assures that the output voltage V_(o) can beachieve at both the low and high line input, subject to the constraintthat the duty ratio is a nonnegative number less than 1.

For example, suppose that V_(inmax)=2V_(inmin). The turns ratio n isthen selected so that at V_(inmax), D₀=0. Thus, n=2Vo/V_(inmax). In thiscase, when the input DC voltage is equal to V_(inmin), the converter 100operates as a full-bridge converter with equal conduction time for theswitching elements in the full bridge, i.e., the switching elements103–106 each have a 50% duty ratio. The duty ratio of the switchingelement 107 is 0 (i.e., the switching element 107 is off during theentire period T). When the input DC voltage is V_(inmax), the switchingelements 105–106 each have a duty ratio equal to 0, while the duty ratioof the switching element 107 is 100% (i.e., the switching element 107turns on during the period T). In these two situations, the voltageacross the filter inductor 115 is equal to zero, and the ripple currentthrough the inductor 115 is equal to zero. When the input voltage V_(in)changes between V_(inmin) and V_(inmax), the duty ratio of the switchingelements 105–106 change between 50% and 0, and the duty ratio of theswitching element 107 is from 0 to 100%.

In the event the input voltage range is greater than 2:1, at the lowerend of the input voltage V_(in), the converter 100 can operate in thefull-bridge converter mode. In this mode, the duty ratio of theswitching element 107 is 0, and deadtime is used to help regulate theoutput voltage V_(o). The switching elements 103–106 therefore haveequal duty ratios less than 50%. The switching elements 103 and 106 turnon and off simultaneously. The switching elements 104–105 also turn onand off simultaneously, and do not turn on when the switching elements103 and 106 are turned on.

At the upper end of the input voltage V_(in), the converter 100 operatesas described above for the high input voltage condition. In this mode,the control signals for the switching elements are as depicted in FIG.2, with the switching element 107 having a duty ratio less than 100%. Tomaximize the no deadtime operation range, the transformer 110 may beconfigured so that at the high input voltage, the duty ratio of theswitching element 107 is 100%. In this case, the converter 100 loses itsno deadtime operation when V_(in) is less than V_(inmax)/2.Alternatively, if it is desired to operate the converter 100 at a higherinput voltage V_(in), then another mode of operation is to always keepthe switching element 107 on, the switching element 105 off, and theswitching element 106 off. In this case, the converter 100 operates likea half bridge converter. The switching elements 103–104 have equal dutyratios less than 50%, and are never turned on simultaneously. When theswitching elements 103–104 are both turned off, there is no voltageapplied to the primary winding 111, and there is deadtime. When theswitching element 103 is turned on, the voltage on the primary windingV₁₁₁ is equal to half the input voltage, V_(in)/2. When the switchingelement 104 is turned on, the voltage on the primary winding V₁₁₁ isnegative half the input voltage, −V_(in)/2. To further increase theinput voltage range, it is possible to combine the two deadtimeoperational modes described above.

The output filter inductance value of a conventional full-bridgeconverter is determined by the condition that under light load (usually,approximately 5–10% of the full load current), the current through theinductor is kept continuous. Given the same specifications of aconventional full-bridge converter, a comparison can be made between theinductance sizes in the converter 100 and in the conventionalfull-bridge converter. Specifically, let the parameters for the fullbridge converter be denoted by the subscript F. Further, assume the samedesign specifications for the two converters, i.e., input voltage rangeV_(inmax):V_(inmin)=2:1, output voltage V_(o), output current I_(o),switching frequency f and period T (on output rectification waveformsf₀=2f, T₀=2T), duty ratio D (i.e., the primary side of the transformer),D₀=2D (i.e., the secondary side of the transformer), and the turns ratioof the output winding to the input winding=n:1. It should be appreciatedthat the below analysis is based on idealized converter components.

For the full-bridge converter,V_(0,F)=nD_(0,F)V_(in).  (3)

The peak-to-peak current on L_(F) Δi_(L,F) for 0<t<D_(0,F) satisfies$\begin{matrix}{{V_{{rect},F} - V_{0}} = {L_{F}\frac{\Delta\; i_{p,F}}{D_{0}T_{0}}}} & (4)\end{matrix}$

andD _(min,F) =V _(o)/(nV _(in,max))=V _(o)/(2nV _(in,min))=0.5.  (5)

Thus, $\begin{matrix}{{{\Delta\; i_{p,F}} = {\frac{n\; T_{0}V_{{in},\min}}{L_{F}}\frac{V_{i} - V_{{in},\min}}{V_{in}}}},} & (6)\end{matrix}$ Δi _(p,F,max)=0.5T ₀ V _(o) /L _(F)(7)

-   -   For the converter 100, $\begin{matrix}        {{V_{o} = {\frac{{nV}_{in}}{2}( {1 + D_{0}} )}},} & (8)        \end{matrix}$        and the relation between the voltage across and the current        through the inductor 115 is $\begin{matrix}        {V_{115} = {{V_{rect} - V_{o}} = {{L_{115}\frac{\mathbb{d}i}{\mathbb{d}t}} = {L_{115}{\frac{\Delta\; i_{L_{115}}}{D_{0}T_{0}}.}}}}} & (9)        \end{matrix}$

So, the peak-to-peak inductor L₁₁₅ current is $\begin{matrix}{{{\Delta\; i_{L_{115}}} = {{\frac{D_{0},T_{0}}{L_{115}}( {{nV}_{in} - V_{0}} )} = {\frac{{nT}_{0}}{L_{115}}\frac{1}{V_{in}}( {V_{{in},\max} - V_{in}} )( {V_{in} - V_{{in},\min}} )}}},} & (10)\end{matrix}$

which has a maximum value $\begin{matrix}{{\Delta\; i_{L_{115}\max}} = {( {\sqrt{2} - 1} )^{2}\frac{T_{0}}{L_{115}}{V_{o}.}}} & (11)\end{matrix}$

From equations (7) and (11) above for peak-to-peak currents, if the fullbridge converter and the converter 100 have the same inductance valueL₁₁₅=L_(F)=L, then $\begin{matrix}{\frac{\Delta\; i_{L,F,\max}}{\Delta\; i_{L_{115},\max}} = {\frac{0.5}{( {\sqrt{2} - 1} )^{2}} = {2.914.}}} & (12)\end{matrix}$

So, when using the same inductor in the output filters, the peak-to-peakcurrent of the converter 100 is only about one-third of that of theconventional full-bridge converter. If the two converters have the samepeak-to-peak current value,Δi_(L,F,max)=Δi_(L) ₁₁₅ _(,max),  (13)

thenL_(F)=2.914L₁₁₅.  (14)

In this case, the inductance of the converter 100 is nearly one-third ofthat of conventional full-bridge converter. It can then be expected thatthe inductor current i_(L115) has a slew rate approximately three timesfaster than that of the conventional full bridge converter. Further, themaximum peak-to-peak voltage drop across the inductor L₁₁₅ decreases incomparison to the full-bridge converter. So, the physical size of theinductor core of the converter 100 can be smaller than the size of thecore used for the filter inductor of the conventional full-bridgeconverter.

FIG. 3 a depicts an alternative embodiment 300 of the DC-DC converter100 of FIG. 1. It is noted that the ideal switching elements 103–107 ofthe converter 100 are replaced in the DC-DC converter 300 by non-idealMOSFET switching elements 303–307, respectively. Each of the MOSFETs303–307 is depicted in FIG. 3 a with its inherent body diode to indicatethe direction of its current blocking capability when turned off. It isunderstood that any other suitable components may be employed for theswitching elements 303–307. The DC-DC converter 300 is configured sothat the switching element 107 can be controlled bi-directionally. Asshown in FIG. 3 a, two MOSFETs 301–302 are used to implement thefunction of the switching element 107. Thus, the switching element 107is both bi-directional current carrying and bi-directional blocking. Theswitching elements 303–306 operate at the same frequency. The switchingelement 303 is off when the switching element 305 is on (see thewaveforms for control signals V₁ and V₃, FIG. 3 b). Further, theswitching element 304 is off when the switching element 306 is on (seethe waveforms for control signals V₂ and V₄, FIG. 3 b). Moreover, thetime sequences of the other control signals are essentially the same asdescribed above with reference to FIG. 1. Specifically, when the controlsignals (i.e., the gate to source voltages) V₆ and V₇ are both high, theswitching elements 301 and 302 both conduct. This permits current toflow between nodes 118 and 119. If either V₆ or V₇ is a low voltagelevel, using the timing signals depicted in FIG. 3 b, then no currentflows between nodes 118 and 119. Therefore, in this illustrativeembodiment, V₅ of FIG. 2 is mathematically represented by logicallyANDing the control signals V₆ with V₇ of FIG. 3 b. As described abovewith reference to FIG. 1, when V₁ is high, the switching element 303 isturned on, and when V₁ is low, the switching element 303 is turned off.In a similar manner, V₂ controls the switching element 304, V₃ controlsthe switching element 305, and V₄ controls the switching element 306.

MOSFETs may be used to form the synchronous rectifier instead of diodes.In this case, self-driven synchronous rectification can simplify thedesign and improve power efficiency because the waveforms on thetransformer windings have no deadtime.

Practical implementations of the presently disclosed DC-DC convertergenerally have waveforms that depart slightly from the idealizedwaveforms described above. For example, transient times and overshootstypically occur at the switching element control voltage transitions.Further, rectifier voltage drops typically cause the output voltagelevels and the voltage levels at the various nodes to change somewhatfrom the levels described above, and power efficiency is slightlyreduced. However, despite these non-ideal results, the operation of theconverter 300 (see FIG. 3 a) is essentially the same as that of theconverter 100 (see FIG. 1).

If the values of the capacitors 108–109 are selected so that there isnon-equal voltage across them, then it is still possible to operate theconverter 300. Specifically, the primary transformer winding 311 hasapproximately zero average voltage across it at each completed switchingcycle, or at the end of any other suitable time period. This may beachieved by non-symmetric operation of the converter 300.

For example, with reference to the idealized converter 100 (see FIG. 1),if the capacitor 108 has a higher voltage than the capacitor 109, thenthe voltage at the node 118, V₁₁₈, becomes less than V_(in)/2. When theswitching elements 103 and 106 are turned on, the voltage across theprimary winding V₁₁₁ is approximately equal to V_(in). Then, theswitching element 106 turns off, and the switching element 107 turns on.The voltage across the primary winding 111 changes from V_(in) toV_(in)−V₁₁₈>V_(in)/2. In the negative cycle, the switching elements 103and 107 turn off. The primary winding voltage V₁₁₁ quickly transitionsto a voltage approximately equal to −V_(in). Then, the switching element108 turns off, and the switching element 107 turns on. The primarywinding voltage V₁₁₁ becomes approximately equal to −V₁₁₈>−V_(in)/2. Inorder to keep volt-second balance on the transformer 110, the averagepositive voltage on the transformer 110 should be approximately equal tothe absolute value of the average voltage on the transformer 110 in thenegative cycle. This can be achieved by altering the duty ratio of atleast some of the switching elements 103–107. For example, the switchingelement 107 may turn on longer in the positive cycle than in thenegative cycle. Alternatively, the switching element 103 may have ashorter duty ratio than the switching element 102. Deadtime could alsobe used to create non-symmetric operation. Other methods are alsopossible. When capacitors are used as the electrical elements 108–109, apractical constraint is that the duty ratios should be selected to givethe capacitors 108–109 sufficient time to recharge.

When batteries or other DC-DC converters are used for the electricalelements 108–109, the voltages across the elements 108–109 areessentially constant. In this case, it is not necessary to recharge theelectrical elements 108–109. Further, it would not be necessary to applyan input voltage across the input terminals 101–102. For example, insolar arrays, solar cells are typically strung together in series. Suchsolar cells may be employed for the electrical elements 108–109.Likewise, batteries are frequently strung together. The seriescombination of such batteries may therefore be taken as the electricalelements 108–109. It should be appreciated that any other suitableelectrical elements may be employed. The switching elements 103–106would be operated so that the average voltage across the transformer110, over an adequate length of time, is approximately equal to zero.

FIG. 4 depicts a second alternative embodiment 400 of the DC-DCconverter 100 of FIG. 1. In the illustrated embodiment, the DC-DCconverter 400 has the Zero Voltage Switching (ZVS) property. Althoughthe control signals V₁–V₄ and V₆–V₇ of the converter 400 are not phaseshift signals, the ZVS property can be obtained via the proper timeselection of the control signals V₁–V₄ and V₆–V₇. The ZVS property forthe switching elements 301–302 is realized independently of loadcondition, whereas for the other switching elements 303–306, it isdependent on the load condition and the circuit parameters (as isgenerally the case for ZVS realization).

All of the capacitors in parallel with the switching elements 301–306 ofFIG. 4 are the output capacitance, C_(oss), of the respectivecapacitors. The inductor 422, L_(R), utilized as a resonant inductancein the transient process, may be the leakage inductance of thetransformer 110. The inductor 422 may alternatively be an externalseries inductance added to broaden the ZVS range. The inductor 422 is inseries with the primary winding 111, and is shown connected to a firstconnection of the primary winding 111, but could be connected to thesecond connection of the primary winding instead.

With reference to FIG. 5, the time sequence from time t₁ to t₉ is oneoperation cycle. Consider the time interval t₁<t<t₂. With the turn offof switching element 306 at t₁, the current through L_(R), which cannotchange instantly, begins to charge C₃₀₃ (see FIG. 4). This makes thedrain-source voltage of the switching element 306, V_(DS,306)=V₁₁₉−V₁₀₂,increase. In the meantime, C₃₀₅ and C₃₀₂ discharge, and the drain-sourcevoltage of the switching element 305, V_(DS,305)=V₁₀₁−V₁₁₉, and thedrain-source voltage of the switching element 302, V_(DS,302)=V₁₁₈−V₄₂₁,decrease. Note that the drain-source voltage of the switching element301, V_(DS,301)=V₁₁₉−V₄₂₁, is zero because it is kept conducting duringthis period. When the drain voltage V₁₁₉ of the switching element 306increases to V_(in)/2, the body diode of the switching element 302conducts, and its drain-source voltage V_(DS,302) is clamped to zero.Then, at t₂, the switching element 302 is driven on at zero voltage bythe control circuit, and takes over the primary current by shortcircuiting its body diode. The loss-less ZVS of the switching element302 is realized. The primary voltage during this time changes fromV_(in) to V_(in)/2. The energy keeps transmitting to the output loadwith decreased power.

Consider the time interval t₂<t<t₃. Both of the switching elements301–302 are conducting. This is a power conversion stage. The primarycurrent is approximately equal to the reflected secondary inductorcurrent, which is decreasing during this period.

Consider the time interval t₃<t<t₄. At time t₃, the beginning of thistransition interval, the switching elements 303 and 301 are turned offby the control circuit. C₃₀₃, C₃₀₆ and C₃₀₁ are charged, and theirdrain-source voltages increase. Correspondingly, C₃₀₄ and C₃₀₅discharge, when the voltage between the nodes 120 and 119 changes fromV_(in)/2 to zero, then to negative. The secondary side rectifierscommute during this period. The current through LR continues to drivethe charging and discharging of the capacitors as described above. Afterthe drain-source voltages of the switching elements 304–305 drop tozero, their source voltages are clamped to ground and V_(in),respectively. Then, the two switches are turned on at zero drain-sourcevoltage by the control circuit.

Consider the time interval t₄<t<t₅. At time t₄, the switching elements304–305 are on. The voltage across the primary winding is V_(in), andthe current in the primary winding changes to the opposite direction.Power conversion continues at this stage with more power transmitted tooutput load.

The operation of the converter 400 during the time intervals t₅<t<t₆,t₆<t<t₇, t₇<t<t₈ and t₈<t<t₉ is similar to the operation during theintervals described above, with the exception that the charging and thedischarging of the capacitors are reversed. It is noted that theconverter 400 with the ZVS property can utilize the no deadtimecharacteristic to easily accomplish self-driven synchronousrectification at the output.

FIG. 6 depicts a third alternative embodiment 600 of the converter 100(see FIG. 1) in a push-pull configuration. The DC-DC converter 600comprises a push-pull converter including switching elements 603–604,and primary windings 613–614 of a transformer 621. The primary winding613 and switching element 603 are connected in series across inputterminals 601–602 to form a node 618. Further, the switching element 604and the primary winding 614 are connected in series across the inputterminals 601–602 to form a node 619. Moreover, electrical elements611–612 are connected in series across the input terminals 601–602 toform a node 620. It is noted that the electrical elements 611–612 aredepicted as capacitors in the presently disclosed embodiment. It isappreciated, however, that other suitable electrical elements, e.g.,batteries, other DC-DC converters, or combinations thereof, may beemployed. A switching element 605 connects the nodes 620 and 618, andprovides a one-direction current path from the node 618 to the node 620.A switching element 606 connects the nodes 620 and 619, and provides aone-direction current path from the node 620 to the node 619. Theswitching elements 603–604 are configured as respective MOSFETs,however, any other suitable electrical element may be employed. Theswitching elements 605–606 may be implemented by any suitable electricalelements that perform one-direction switching. As shown in FIG. 6, theswitching element 605 includes two MOSFETs 607–608 in series, and theswitching element 606 includes two MOSFETs 609–610 in series. Any othersuitable electrical components, e.g., one MOSFET and one diode connectedin series to perform one-direction switching, may be employed in placeof the serially connected MOSFETs. In the illustrated embodiment, thevoltage across the capacitor 611 is equal to the voltage across thecapacitor 612. Accordingly, the voltage at the node 620 is approximatelyequal to one half of the input voltage V_(in).

The voltage across a center-tapped secondary winding 615 is applied to arectifier 616 to obtain a rectified voltage V_(rect), which in turn isapplied to a filter circuit 617 (e.g., a low pass filter). The outputvoltage is the filtered voltage taken across nodes 625–626.

FIG. 7 depicts illustrative waveforms of the DC-DC converter 600 (seeFIG. 6) including idealized components. It should be appreciated thatthe time sequences of the control signals V₁₃, V₁₄, V₁₅ and V₁₆ in FIG.7 are described herein for purposes of illustration, and that othersuitable time sequences are possible. High voltage levels applied to thecontrol signals of the switching elements cause the switching elementsto close, and low voltage levels cause the switching elements to open.V₁₁, V₁₂, V₁₃, V₁₄, V₁₅, and V₁₆ are the control signals of theswitching elements 603, 604, 607, 608, 609, and 610, respectively.Suppose the converter 600 works in steady state, and its output inductorcurrent is under continuous conduction mode. The dot ends of thewindings of the transformer in FIG. 6 refer to as positive.

From time t₀ to t₁, the switching element 603 is turned on. The voltagesV₆₁₃ and V₆₁₄ of the transformer primary windings 613–614 areapproximately equal to the input voltage V_(in). During this period, theinput current i_(in) increases, and equals the current i₆₁₃ (which ispositive from the terminal 601 to the node 618) of the primary winding613, and reaches to I_(p,max) at time t₁.

At time t₁, the switching element 603 is off, and the switching element605 (i.e., the MOSFETs 607–608) is on. The primary winding voltages V₆₁₃and V₆₁₄ are approximately equal to V_(in)/2, and the input currenti_(in) decreases from i_(p,max)/2. Also, i₆₁₃, decreasing fromi_(p,max), is supplied by i_(in), the discharging current of thecapacitor 611 and the charging current of capacitor 612. The currenti₆₁₄ remains zero.

At time t₂, the switching element 605 (i.e., the MOSFETs 607–608) turnsoff. The switching element 604 turns on. After a short transient time,the voltages V₆₁₈ and V₆₁₉ of the primary windings 618 and 619,respectively, become approximately equal to the negative input voltage,−V_(in). The primary winding current i₆₁₉ (which is positive from thenode 619 to the terminal 602) of the primary winding 619 transits from avalue of zero immediately before t₂, to a value of i_(p,min) just aftert₂. Meanwhile, i₆₁₃ goes from I_(p,min) to zero. Then, from t₂ to t₃,i₆₁₄ changes from i_(p,min) to i_(p,max).

At time t₃, the switching element 604 turns off, and the switchingelement 606 (i.e., the MOSFETs 609–610) turns on. Then, the primarywinding voltages V₆₁₃ and V₆₁₄ are approximately equal to −V_(in)/2. Theinput current i_(in) becomes approximately equal to i_(p,max)/2. Aftert₃, the primary winding current i₆₁₄ changes from i_(p,max) towardsi_(p,min), reaching i_(p,min) at t₄. From t₃ to t₄, the primary windingcurrent i₆₁₄ is supplied by the input current i_(in), the chargingcurrent of capacitor 611, and the discharging current of capacitor 612.The current i₆₁₃ remains zero. From time t₄, after a very shorttransient time, V₆₁₃ and V₆₁₄ change polarity and the process repeats asdescribed above.

When the converter 600 (see FIG. 6) operates as described above, it isin the no deadtime operation mode. It is noted that the output voltageequations, and the relationship of the output filter inductance to theripple current passing through the output inductor, for the converter600 are essentially the same as those described above with reference tothe converter 100 (see FIG. 1).

The voltage V_(rect) is the absolute value of the primary windingvoltage V₆₁₃ times the transformer turns ratio n. The filtered (i.e.,averaged) value of V_(rect) is the output voltage V_(o). It is possibleto regulate the output voltage V_(o) by controlling the amount of timethat V_(rect) has value V_(in) versus value V_(in)/2. Because V_(rect)in FIG. 6 is like V_(rect) in FIG. 3 a, benefits of the proposedtopology, in comparison with conventional push-pull converters include(1) reduced inductor volume, (2) reduced inductance value, and (3)reduced input current-ripple. Further, it is possible to introducedeadtime to regulate the output voltage when input voltage range isgreater than 2:1.

By operating the switching elements as described above, the capacitorscan charge and discharge on each cycle. Although different operationaltiming of the switching elements is possible, when capacitors are usedfor the switching elements, it is important to keep charge balance onthe capacitors to maintain constant voltage across the capacitors. Ifbatteries or other suitable electrical elements are employed, this isnot a concern.

FIG. 8 depicts a fourth alternative embodiment 800 of the converter 100(see FIG. 1) in a dual-forward implementation. The DC-DC converter 800operates in essentially the same manner as the DC-DC converter 600, withthe exception that there are two transformers, each with a primary andsecondary winding. Further, a transformer reset is employed to maintainflux balance in the core.

FIGS. 9 a–9 b depict illustrative electrical components and connectionsthat may be employed in the DC-DC converters 100, 300, 400, 600, and800. As shown in FIGS. 9 a–9 b, there are two electrical elements904–905 connected in series across two input terminals 901 and 902. Anode 903 between the electrical elements 904–905 has a voltage,V_(in)/2. There is also a primary winding 911. In FIG. 9 a, the primarywinding 911 has one of its connections coupled to two switching elements906 and 909. The switching element 909 connects a node 910 to the inputterminal 902. The switching element 906 (shown as the serially connectedMOSFETs 907–908) connects the node 903 to the node 910. FIG. 9 b depictssimilar connections, with the exception that the primary winding 911 isin series with a resonant inductor 912, for application in convertershaving the ZVS property.

In the illustrated embodiment, the switching elements 906 and 909 arenot on at the same time. When the switching element 906 is on andswitching element 909 is off, the voltage at the node 910 is equal tothe voltage at the node 903, i.e., V_(in)/2. When the switching element906 is turned off and the switching element 909 is turned on, thevoltage at the node 910 is equal to the voltage at the node 902, whichfor purposes of illustration is assumed to be ground. Thus, the primarywinding 911 has one end connected to the electrical element 905 atV_(in)/2 volts, or connected to ground at 0 volts. It is also possibleto have both of the switching elements 906 and 909 turned off.

In the event the second connection of the primary winding 911 isconnected to the input terminal 901, perhaps via one or more switchingelements, then Vleg of FIGS. 9 a–9 b may be V_(in) or V_(in)/2,depending on the control signals of the switching elements. Thus, nodeadtime topologies are possible by creating V_(rect) as in FIG. 2.Further, V_(rect) switches from nV_(in) to nV_(in)/2. In conventionaldual-ended converters, the rectified secondary winding voltage typicallyswitches from nV_(in) to 0 volts.

The above description, and the description that follows, are based onideal operation of the DC-DC converters. If capacitors are used as theelectrical elements 904–905, then after a capacitor is discharged, it issubsequently recharged to keep the DC link voltage V₉₀₃ constant. Forexample, if symmetric operation of a converter is utilized, this wouldmean that the charge time and the discharge time of each capacitor arethe same. This is why in FIG. 6 (see also FIG. 8), the switching element604 and the primary winding 614 are connected as shown. In aconventional push-pull converter, it is often desirable to connect thesources of the primary switching elements (which in FIG. 6 would be theswitching elements 603 and 604). This makes driving the switchingelements 603–604 easier because they share the same ground. That is,conventional push-pull configurations would typically connect the firstconnection of the second primary winding 614 to the first input terminal601. The switching element 604 would be connected in series with theprimary winding 614, so that one connection of the switching element 604is connected to the second connection of the second primary winding 615,and the second connection of switching element 604 is connected to thesecond input terminal 602. This configuration is possible for theconverter 600 of FIG. 6 (and the converter 800 of FIG. 8) if theelectrical elements do not require equal charge and discharge, e.g., ifthe electrical elements are batteries or other voltage sources. On theother hand, if the electrical elements are capacitors, then additionalcharge and discharge techniques must be used to supplement the circuit.This is because the capacitor 611 would normally not be recharging.However, the DC-link charge balance voltage problem for the push-pullconverter 600 can be corrected by changing the switching elementconfiguration, as depicted in FIG. 10.

The DC-DC converter 1000 (see FIG. 10) comprises the cascade connectionof a push-pull DC-DC converter 1022 with a three-level switch cell 1021.The push-pull converter 1022 includes one transformer with two primarywindings 1012–1013. A switching element 1005 is in series with the firstprimary winding 1012, and is connected across input terminals 1019–1020.Similarly, a switching element 1006 is in series with the second primarywinding 1013, and is connected across the nodes 1019–1020. In thisconfiguration, it is possible, although not necessary, to have thesource of the switching elements 1005–1006 connected, as depicted inFIG. 10. As described above, this makes it simpler to control the turnon and the turn off of these switching elements. As shown in FIG. 10, asecondary winding 1014 is center tapped and connected to a rectifier1015. It is understood that other suitable secondary winding andrectifier configurations are possible. The rectified voltage V_(rect) isapplied to a filter 1016, and the output voltage is taken acrossterminals 1017–1018.

The three-level switch cell 1021 includes the two input terminals1001–1002 and two output terminals 1019–1020. The two output terminalsof the three-level switch cell 1021 are the two inputs of the push-pullDC-DC converter 1022. Electrical elements 1009–1010 are connected inseries across the input terminals 1001 and 1002, forming a DC-link node1011, which has a voltage of half the input voltage, V_(in)/2. Aswitching element 1003 connects the three-level switch cell inputterminal 1001 to the node 1019, which is a first input to the push-pullcircuit 1022. Another switching element 1004 connects the three-levelswitch cell second input terminal 1002 to the node 1020. Diodes areemployed as switching elements 1007–1008, which are connected in seriesacross the nodes 1019–1020. As described above, other suitable types ofswitching elements could be used, provided they have reverse currentblocking capabilities. A node 1011 between the two diodes 1007–1008 isthe DC-link node.

The converter 1000 operates with no deadtime provided that the inputvoltage range is within 2:1. Otherwise, deadtime is used for regulationof the output voltage. For purposes of illustration, it is assumed thatthe input voltage range is less than 2:1. FIG. 11 depicts illustrativecircuit waveforms for the converter 1000. A signal S₁ controls theswitching element 1003. When S₁ is a high voltage level, the switchingelement 1003 is turned on. When S₁ is a low voltage level, the switchingelement 1003 is turned off. Similarly, a signal S₂ controls theswitching element 1004, a signal S₃ controls the switching element 1005,and a signal S₄ controls the switching element 1006.

As shown in FIG. 11, the switching elements 1005–1006 are turned oncomplimentarily with 50% duty cycle. The switching elements 1003–1004have the same duty ratio, and operate at the same switching frequency asthe switching elements 1005–1006. Regulation of the output voltage isachieved by adjusting the duty ratios of the switching elements1003–1004. When S₁ and S₂ are both high, the switching elements1003–1004 are both turned on. Full input voltage is applied to thepush-pull converter 1022 because V_(bus)=V_(in). Assume that S₄ is high,also, and therefore S₃ is low. Then, the switching element 1006 is on,and the switching element 1005 is off. Full input voltage is applied tothe second primary winding 1013, and the input current equals the secondprimary winding current I₁₀₁₃. No current passes through the diodes 1007and 1008. After this, S₂ goes to a low voltage level. Further, S₁ and S₄remain high, while S₃ remains low. The switching element 1004 turns offand blocks current (the body diode of the MOSFET 1004 also blockscurrent). Current is diverted through the diode 1008. The DC linkvoltage at the node 1011 is V_(in)/2, and therefore the voltage at thenode 1020 becomes ideally V_(in)/2. Hence, after a short transitiontime, V₁₀₂₀ changes from 0 volts to V_(in)/2 when the switching element1004 turns off, and subsequently, V_(bus) changes from V_(in) toV_(in)/2. The primary winding current i₁₀₁₃ is supplied by thedischarging capacitor current i₁₀₀₉, and by the charging capacitorcurrent i₁₀₁₀. The input current is equal to charging capacitor currenti₁₀₁₀.

The next half cycle of operation is symmetric to that described above.From S₁, S₄ high and S₂, S₃ low, the next state is to turn S₄ low, turnS₃ high, and turn S₂ high. S₁ remains high. Because both S₁ and S₂ arehigh, the switching elements 1003 and 1004 are on. Switching elements(i.e., the diodes) 1007–1008 are off, and V_(bus) is equal to the inputvoltage V_(in). Current flows through the first primary winding 1012from the node 1019 into the switching element 1005, and is equal to theinput current, i_(in). No current flows through the primary winding 1013(after a short transition time). In the next state, S₁ turns off. Theswitching element 1003 turns off and blocks current (the body diode ofthe MOSFET 1003 also blocks current). Current is diverted through thediode 1007 from the input node 1001 through the electrical element 1009.The diode 1008 blocks current, and thus the switching element 1008 isoff. The DC link voltage at the node 1011 is V_(in)/2, and therefore thevoltage at the node 1019 ideally becomes V_(in)/2. Hence, after a shorttransition, V₁₀₁₉ changes from V_(in) volts to V_(in)/2 after theswitching element 1003 turns off. The voltage at the node 1020 remainsat approximately zero volts. Therefore, V_(bus) changes from V_(in) toV_(in)/2 when the switching element 1003 turns off. The primary windingcurrent i₁₀₁₂ is supplied by the charging capacitor current i₁₀₀₉, andby the discharging capacitor current i₁₀₁₀. The input current is equalto the charging capacitor current i₁₀₀₉.

By utilizing the symmetric operation above, the capacitors have equalcharge and discharge times. Thus, the DC-link voltage at the node 1011can be kept at V_(in)/2. Further, by creating V_(bus), as describedabove and as depicted in FIG. 11, no deadtime operation is achieved. Forthe push-pull winding configuration shown, V_(rect)=nV_(bus), in which nis the number of secondary winding turns divided by the number ofprimary winding turns. The output voltage is the filtered (i.e.,averaged) value of V_(rect). As in the other topologies described above,regulation of output voltage is achieved by controlling the length oftime that V_(rect) is equal to V_(in)/2, and by controlling the lengthof time that V_(rect) is equal to V_(in). Thus, the advantages of thepreviously described converters 100, 300, 400, 600 and 800 are achieved.The lower output filter inductance value and volume are required becauseV_(rect) has a minimum value of V_(in)/2, and not necessarily 0 volts.

As in the other circuit topologies described above, in the event theinput voltage range is greater than 2:1, deadtime can be used toregulate the voltage. For example, when the input voltage is high, bothof the switching elements 1003–1004 can be turned off to provide aV_(bus) level equal to zero volts. Therefore, V_(rect) would also bezero volts. There are several ways to implement this deadtimeeffectively. For example, assume the switching elements 1003–1004 arealternately turned on with equal time and a duty ratio less than 50%.Further, let the switching elements 1005–1006 be turned oncomplimentarily with 50% duty cycle. Moreover, restrict S₁ and S₃ totransition to high at the same instant of time, and restrict S₂ and S₄to transition to high at the same instant of time. Then, when either S₁or S₂ are high, V_(bus)=V_(in)/2. When both S₁ and S₂ are off,V_(bus)=0.

A common feature of the DC-DC converter of FIG. 9 a and the DC-DCconverter 1000 (see FIG. 10) is that the converter 1000 includes theelectrical circuit of FIG. 9 a, only with different types of switchingelements. That is, comparing FIG. 9 a to FIG. 10, (1) the inputterminals 901–902 correspond to the input terminals 1001–1002, (2) theelectrical elements 1009–1010 correspond to the electrical elements904–905, (3) the switching element 906 corresponds to the switchingelement 1007, (4) the switching element 909 corresponds to the switchingelement 1003, and (5) the primary winding 911 corresponds to the primarywinding 1012. The types of switching elements employed depends on thespecific directional conducting and blocking requirements of theswitching elements. However, from the point of view of ideal switchingelements, the circuit of FIG. 9 a is included in the converter 1000 ofFIG. 10.

There are benefits to using the three level switching cell 1021 (seeFIG. 10) because it can be used with any DC-DC converter, as shown inFIG. 12. Specifically, FIG. 12 depicts the output terminals 1019–1020 ofthe three level switch cell 1021, which can be connected to the inputterminals of a DC-DC converter. The purpose of the three level switchcell 1021 is to create an input voltage of the DC-DC converter that canbe equal to one of three levels—V_(in), V_(in)/2, or 0 volts. Assumingsymmetric operation, the duty ratio of the switching element 1003 equalsthe duty ratio of the switching element 1004. For example, suppose bothof the switching elements 1003 and 1004 are on. Then, V_(bus)=V_(in),and the output current provided by the switching cell into the DC-DCconverter is provided by source input V_(in). Suppose that the switchingelement 1003 turns off while switching element 1004 remains on. Then,the diode (i.e., the switching element) 1007 conducts, and the voltageat the node 1019 becomes the DC-link voltage V₁₀₁₁, andV_(bus)=V_(in)/2. The output current of the three-level switching cell1021 is provided by the charging capacitor current i₁₀₀₉ and thedischarging capacitor current i₁₀₁₀. In order to keep the constantDC-link voltage V₁₀₁₁, if capacitors are used as the electrical elements1009–1010, charge balance on the capacitors must be maintained. One wayto achieve this is as described above with reference to the operation ofconverter 1000. That is, in the next half cycle of operation, theswitching element 1003 turns on while the switching element 1004 remainsoff, in order to create V_(bus)=V_(in)/2. In this mode, the diode (i.e.,the switching element) 1008 conducts current, V₁₀₂₀=V_(in)/2, and thusV_(bus)=V_(in)/2. The output current of the three-level switching cell1021 is provided by the discharging capacitor current i₁₀₀₉ and thecharging capacitor current i₁₀₁₀. By keeping the operation of the threelevel switching cell 1021 symmetric so that the duty ratio of switchingelement 1003 equals the duty ratio of switching element 1004, thecapacitors have equal charge and discharge times. So, the DC-linkvoltage V₁₀₁₁ can be kept approximately constant. Similar to before, ifthe voltages across the capacitors are not equal, then non-symmetricoperation of the switches may be used to maintain charge balance.Finally, there is a third possible state for the switching cell 1021, inwhich both of the switching elements 1003–1004 are turned off. In thiscase, V_(bus)=0, and there is no output current to the three levelswitching cell 1021.

The benefits of using the three level switching cell 1021 vary dependingon the topology employed. For isolated dual-ended DC-DC converters, suchas push-pull and full bridge, it is possible to operate the dual-endedDC-DC converter with primary switches having a 50% duty ratio. Thisreduces the input current ripple. So, when the three level switch cell1021 is used, as described above, the benefits may include (1) nodeadtime operation when the input voltage range is less than 2:1, (2)lower output filter inductance value, and (3) lower output filterinductance volume.

FIGS. 13 a–13 c depict alternative embodiments 1300 a–1300 c of DC-DCconverters including the three level switch cell 1021. As shown in FIGS.13 a–13 c, the converter 1300 a is configured as a full-bridgeconverter, the converter 1300 b is configured as a dual-forwardconverter, and the converter 1300 c is configured as an asymmetricalhalf-bridge converter.

As shown in FIG. 13 a, the three level switch cell 1021 is connected tothe full-bridge converter 1201. Suppose the input voltage range is lessthan 2:1. Then, for this converter, the switching elements 1003–1004operate with same duty ratio, but with 180° phase shift. Switchingelements 1301 and 1304 turn on and off together, and complementary-toswitching elements 1302–1303, which also turn on and off together. Eachswitch has a 50% duty cycle, and the same switching frequency as theswitching elements 1003–1004. When both of the switching elements1003–1004 are turned on while the switching elements 1301 and 1304 areturned on at the same time, a full input voltage is applied to theprimary winding of transformer. Thus, V_(bus)=V_(in). The input currentequals the primary winding current. Next, the switching element 1004 isturned off while the switching element 1003 is kept on. The diode (i.e.,the switching element) 1008 conducts. So, the voltage at the node 1020becomes the DC link voltage, and V_(bus)=V_(in)/2. The output inductoris discharged with half the input voltage. The capacitor 1009discharges, and the capacitor 1010 charges. The primary winding currentis supplied with the discharging current of capacitor 1009 and thecharging current of capacitor 1010. The input current is equal tocharging current of capacitor 1010.

The next half cycle of operation is symmetrical to the first one. Theswitching element 1004 is turned on again, so that both of the switchingelements 1004 and 1003 are turned on. At the same time, the switchingelements 1302–1303 are turned on, and switching elements 1301 and 1304are turned off. Full input voltage is applied to charge the outputinductor, and V_(bus)=V_(in). The input current equals the primarywinding current. Next, the switching element 1003 is turned off whilethe switching element 1004 is kept on. Further, the diode (i.e., theswitching element) 1007 conducts. So, the voltage at the node 1019becomes the DC-link voltage, and V_(bus)=V_(in)/2. The output inductoris discharged with half the input voltage. The capacitor 1009 ischarged, and capacitor 1010 is discharged. The primary winding currentis supplied with the charging current of the capacitor 1009, and thedischarging current of the capacitor 1010. The input current is equal tocharging current of the capacitor 1009.

When operated as described above, V_(rect)=nV_(bus), in which n is thetransformer turns ratio. Thus, V_(rect) and V_(bus) are as describedabove with reference to the converter 1000 (see FIG. 10). Therefore, theinductance value and the inductor volume of the output filter isreduced.

For wider input voltage range, it is possible to apply deadtime forcontrol purposes, or to make V_(bus)=0. This is also possible for inputvoltage ranges of less than 2:1. Any combination of V_(bus) levels arepossible as the input of the DC-DC converter 1201 (see FIG. 12).However, charge balance issues are to be dealt with if capacitors areemployed, as well as keeping a zero average voltage value across thetransformer.

FIGS. 13 b–13 c depict other dual-ended topologies that employ thethree-level switch cell 1021. It is understood that other suitabletopologies are also possible. The operation is similar to that of theconverters 1000 and 1300 described above, and similar benefits areachieved.

The benefits of reduced inductor size using the three level switch-cell1021 can also be achieved in isolated single ended converters, such asthe forward converter 1400 of FIG. 14. In the converter 1400, the outputterminals of three-level switch cell 1021 are connected to the inputterminals of a single ended forward converter. This is just one type ofsingle ended topology that may be used, and other suitable single endedtopologies are possible. A reset winding for the forward converter isnot shown. Alternatively, clamp or other techniques may be employed tomaintain flux balance on the core. The technique to achieve this is notimportant for understanding the basic principles of operation of theconverter 1400.

As shown in FIG. 15, V_(bus) is illustrated as T periodic having a valueof either V_(in), V_(in)/2, or 0 volts. Suppose the switching elements1003–1004, and the switching element 1402 are turned on at the sametime. Full input voltage is applied to the primary winding 1403 of thetransformer, making V_(bus)=V_(in). The inductor current i₁₄₀₈increases. Then, the switching element 1004 is turned off while theswitching elements 1003 and 1402 remain on. The diode (i.e., theswitching element) 1008 turns on. So, the voltage at the node 1020becomes the DC link voltage, and V_(bus)=V_(in)/2. The inductor currentdecreases because V_(rect)=nV_(in)/2<V_(o). The capacitor (i.e., theelectrical element) 1009 discharges, and the capacitor 1010 charges. Theprimary winding current is supplied with the discharging current of thecapacitor 1009, and the charging current of the capacitor 1010. Theinput current is equal to the charging current of the capacitor 1010.Next, the switching element 1402 is turned off at the same time thetransformer resets. If the switching element 1003 is turned off, as inFIG. 15, then V_(bus)=0. It is noted that the value of V_(bus) isunimportant during the transformer reset, so other values arepermissible. The input current and the primary winding current are zero.The inductor current i₁₄₀₈ flows though the freewheeling diode on thesecondary side, and decreases. The next half cycle of operation issymmetrical to the first half cycle.

The output voltage of the converter 1400 is the filtered (i.e.,averaged) voltage of V_(rect). Therefore, the voltage transfer ratio ofthe converter 1400 is $\begin{matrix}{\frac{V_{o}}{V_{i}} = {{n( {D_{1} + \frac{D_{2}}{2}} )}.}} & (15)\end{matrix}$Here, the duty ratio D1 is the fraction of each period thatV_(bus)=V_(in), and the duty ratio D2 is the fraction of each periodthat V_(bus)=V_(in)/2, as indicated in FIG. 15. It is noted that n isthe turns ratio of the transformer, as described above. The inductorcurrent ripple is smaller for the converter 1400 compared to theconventional single ended forward converter 1401. This is shown bycalculating the volt-second product across the inductor 1408, and byshowing that it is smaller than that of the conventional 2-levelconverter 1401.

The volt-second product of the converter 1400: $\begin{matrix}{\frac{V_{o}{T( {1 - D_{1} - {D_{2}/2}} )}}{1 + {{D_{2}/2}D_{1}}}.} & (16)\end{matrix}$

The volt-second product of the conventional converter 1401:V_(o)T(1−D).  (17)

It is noted that (1−D)=(1−D₁−D₂/2) for the same V_(in) and V_(o). Thus,the converter 1400 has reduced the volt-second product for the outputinductor 1408 by a factor of $\begin{matrix}{\frac{1}{1 + {{D_{2}/2}D_{1}}} < 1.} & (18)\end{matrix}$This implies that a smaller core and a smaller inductance value arepossible.

FIGS. 16 a–16 c depict the three-level switch cell 1021 connected toseveral types of current fed DC-DC converters 1600 a–1600 c. FIG. 16 adepicts the converter 1600 a configured as a current-fed push-pull buckconverter, FIG. 16 b depicts the converter 1600 b configured as acurrent-fed full-bridge converter, and FIG. 16 c depicts the converter1600 c configured as a current-fed dual-forward converter. It isunderstood that alternative configurations are possible.

DC-DC converters are considered to be voltage fed converters when thefilter inductor is located on the secondary side of the transformer. Incurrent fed topologies, the inductor is moved to the primary side of thetransformer. This has benefits in multi-output DC-DC converters becauseonly one inductor core is required. The three level switch-cell 1021connected to current fed topologies achieves the benefits of reducedvalue and reduced volume of the inductor, which is on the primary side.The voltage stresses on the switching elements are reduced as well.

The operation of current fed topologies is similar to that of voltagefed topologies. For example, the current fed half-bridge is depicted inFIG. 17 with the three level switch cell. The output of three levelswitch cell 1021 is connected to the input of current fed half bridge1701. Because the two capacitors 1704–1705 operate as a voltage source,the elements 1021 and 1704–1706 make up a buck stage.

The operation of DC-DC converter 1700 is similar to that of theconverter 1400. The switching elements 1003–1004 and 1703 are turned onat the same time, and the switching element 1704 is off. The full inputvoltage is applied, and the input voltage to the current fed half bridgeconverter 1701 is V_(bus)=V_(in). In current fed topologies,V_(rect)=V_(o). This output voltage is reflected to the primary winding,and the primary winding voltage is V_(o)/n, in which n is the turnsratio, as defined above. The capacitors 1704–1705 each have a steadystate voltage equal to the reflected output voltage,V₁₇₀₄=V₁₇₀₅=V_(o)/n. So, when the switching element 1703 is on, theinductor has a positive voltage across it, V_(in)−2V_(o)/n, and theinductor current I₁₇₀₆ increases. Next, the switching element 1004 isturned off while the switching elements 1003 and 1703 remain on. Thediode (i.e., the switching element) 1008 conducts. The voltage at thenode 1019 becomes the DC-link voltage. So, the voltage across theinductor becomes V_(in)/2−2V_(o)/n, which is less than zero assumingless than 2:1 input voltage range. Thus, the inductor current i₁₇₀₆decreases. The next half cycle of operation is symmetrical to the firsthalf cycle. Charge balance of the capacitors included in the converter1700 is achieved via symmetrical operation of the converter.

Without the three-level switch cell 1021, the inductor voltage wouldswitch between V_(in)−2V_(o)/n and −2V_(o)/n, which is a larger peak topeak value compared to the value in the converter 1700. Thus, theconverter 1700 requires a reduced inductance value.

For an input voltage range greater than 2:1, the switching elements1003–1004 can both turn off, so that V_(bus)=0 volts. In this case, theinductor current will freewheel through the diodes 1007–1008. It shouldalso be noted that sometimes an additional small inductor is added tothe output filter 1710, for eliminating the transient current shock. Theoperation of the current fed half bridge is typical for the three-levelswitch cell connected to other current fed topologies. It is understoodthat the topologies of FIG. 16 are not exhaustive, and other suitabletopologies are possible.

FIG. 18 depicts the three-level switch cell used in a three-level buckconverter 1800 that has no isolation. Three-level switch cell 1021 isattached directly to an output filter 1803. Thus, the output voltageV_(o) taken across terminals 1801–1802 is the filtered (i.e., averaged)value of V_(bus). The operation of the DC-DC converter 1800 is similarto that of the converter 1700. The switching elements 1003–1004 areturned on at the same time. The full input voltage, V_(bus)=V_(in),isapplied to the filter 1803. The inductor current i₁₈₀₄ increases, and nocurrent passes through the diodes 1007–1008. Then, the switching element1004 is turned off. The diode (i.e., the switching element) 1008 turnson. So, the voltage at the node 1020 becomes the DC link voltage, andV_(bus)=V_(in)/2. Assuming V_(in)/2<Vo, the inductor current decreases.The capacitor (i.e., the electrical element) 1009 discharges, and thecapacitor 1010 charges. The input current is equal to the chargingcurrent of the capacitor 1010. The next half cycle of operation issymmetrical to the first half cycle. Thus, V_(bus) is the same as shownin FIG. 11.

For the case when V_(in)/2>V_(o), the switching cell operates for sometime interval with V_(bus)=0 volts. One way to achieve this is bysymmetrically operating the three-level switch cell 1021 to switchbetween V_(bus)=V_(in)/2and V_(bus)=0 volts, via proper control of theduty ratios. When V_(bus)=0 volts, both of the switching elements1003–1004 are turned off, and the diodes 1007–1008 are bothfreewheeling. Controlling the length of freewheel time helps regulatethe average value of V_(bus) (which is equal to V_(o)). To createV_(bus)=V_(in)/2, the switching elements 1003–1004 are alternatelyturned on an off in a symmetrical manner, as described above. Theconverter 1800 is referred to herein as a three-level switch cell buckconverter.

Three-level switch cell buck converter 1800 can be used in a mannersimilar to the three-level switch cell 1021 in FIG. 12. FIG. 19 showsthat the three-level switch cell buck converter 1800 has an output thatcan be directly connected to the input of another DC-DC converter 1901to form a single DC-DC converter 1900. Although it is possible toseparately regulate the output V_(o) of DC-DC converter 1901 from theoutput V_(Buck) of the three-level switch cell buck converter 1800,there are classes of converters in which DC-DC converter is notseparately controlled. For example, in the case when DC-DC converter1901 is a dual-ended converter, it is possible to operate theDC-transformer switches in 1901, as shown in FIG. 20, at 50% duty ratio.The output voltage of DC-DC converter 1900, Vo, is regulated bycontrolling the output voltage of the three-level switch cell buckconverter 1800. That is, Vo=nV_(Buck). Thus, the duty ratios of thethree-level switch cell buck switching elements are used to control theoutput of the isolated dual-ended DC-DC converter V_(o). Such aconfiguration is generally referred to as a two-stage DC-DC converter,as shown in FIG. 20.

For the two stage converter 2001, it is possible to keep the duty ratioof the DC-DC transformer 2003 (half-bridge, full-bridge, push-pull,dual-forward, etc.) switches to be 50%, which leads to no deadtime, anda smaller inductor 2005 in the output filter 2002. Regulation of outputvoltage V_(o) is obtained by regulating V_(Buck) using the techniquesdescribed above.

FIGS. 21 a–21 g depict specific examples of two-stage DC-DC convertersthat utilize the three-level switch cell buck converter 1800. It isunderstood that these are merely illustrative examples, and othersuitable configurations of such two-stage converters are possible.

Ideally, no inductor is needed in the output filter, but from apractical implementation viewpoint, it is often added to suppress thetransient current and to improve filtering. Hence, the DC-DC converters2100 a–2100 g of FIGS. 21 a–21 g are depicted with and without theoutput filter inductor. When an additional output filter inductor isadded, it is typically small. FIG. 21 a depicts the converter 2100 aconfigured as a three-level two-stage buck and push-pull converter, FIG.21 b depicts the converter 2100 b configured as another embodiment ofthe three-level two-stage buck and push-pull converter, FIG. 21 cdepicts the converter 2100 c configured as a three-level two-stagefull-bridge converter, FIG. 21 d depicts the converter 2100 d configuredas another embodiment of the three-level two-stage full-bridgeconverter, FIG. 21 e depicts the converter 2100 e configured as athree-level two-stage dual-forward converter, FIG. 21 f depicts theconverter 2100 f configured as another embodiment of the three-leveltwo-stage dual-forward converter, and FIG. 21 g depicts the converter2100 g configured as a three-level two-stage half-bridge converter. Itis appreciated that other suitable configurations are possible.

FIGS. 22 a–22 c depict alternative embodiments 2200 a–2200 c of thethree-level switch cell 1021. FIG. 22 a depicts the four-level switchcell 2200 a, which operates like the switch cell 1021, with theexception that it is now possible to create V_(bus)=V_(in),V_(bus)=2V_(in)/3, V_(bus)=V_(in)/3, or V_(bus)=0 volts.

Specifically, suppose that, in the first ⅓ cycle, the switching elements2201 and 2204 are on and the switching elements 2202–2203 are off. ThenV_(bus)=V_(in). Next, the switching element 2204 is turned off, and thediodes 2207–2206 conduct, while the diode 2205 blocks current andremains off. Current flows from the diode 2207 to the diode 2206, andthrough the body diode of the switching element 2202. Thus,V₂₂₁₄=V₂₂₀₉=2V_(in)/3, and therefore V_(bus)=V_(in)/3. Assume that theelectrical elements 2210–2212 are capacitors. Then, the capacitor C₂₂₁₀discharges, while the capacitors C₂₂₁₁ and C₂₂₁₂ charge.

In the second ⅓ cycle, the switching element 2204 is turned on again, sothat both of the switching elements 2201 and 2204 are on. Full voltageis applied, and V_(bus)=V_(in). Next, the switching elements 2201 and2204 are both turned off, and the switching elements 2202–2203 areturned on at the same time. The diodes 2207 and 2205 conduct. One thirdof the input voltage is applied, V_(bus)=V_(in)/3. The capacitors2210–2212 are charged.

In the third ⅓ cycle, the switching elements 2202–2203 are turned offwhile the switching elements 2201 and 2204 are turned on at the sametime. Full voltage is applied, and V_(bus)=V_(in). Next, the switchingelement 2201 turns off. The diodes 2205–2206 conduct, makingV_(bus)=V_(in)/3. The capacitor 2212 discharges, and the capacitors2210–2211 are charged.

Provided the operation as described above is symmetric, the capacitorswill maintain charge balanced. If the electrical elements are notcapacitors, then charge balance is unimportant. Finally, if theswitching elements 2201–2204 are turned off, then V_(bus)=0 volts.

The switch cell 2200 b of FIG. 22 b generalizes the switch cell 2200 aof FIG. 22 a for the case where there are n_(odd) electrical elements,in which n_(odd) is an odd number. The switch cell 2200 c of FIG. 22 ccomprises the generalized (n_(even)+1) level switching cell, in whichn_(even)is an even number.

All of the switching cells 2200 a–2200 c of FIGS. 22 a–22 c can be usedin the same manner as the three-level switching cell 1021 of FIG. 10.For example, they can be directly connected to any DC-DC converter, orthey can be connected to an LC filter to form a general (n+1) levelswitch cell buck converter like the converter 1400 (see FIG. 14). Thisn-level switch cell buck converter can then be connected to any DC-DCconverter like the converters 1900 and 2000 (see FIGS. 19–20).

It will further be appreciated by those of ordinary skill in the artthat modifications to and variations of the above-described DC-DCconverters providing reduced deadtime may be made without departing fromthe inventive concepts disclosed herein. Accordingly, the inventionshould not be viewed as limited except as by the scope and spirit of theappended claims.

1. A DC-DC power converter, comprising: first and second inputterminals; at least one electrical element connected to at least one ofthe first and second input terminals, the at least one electricalelement being operative, in the event a first voltage is applied acrossthe first and second input terminals, to provide a second voltage havinga value between the first voltage value and a reference voltage value;at least one transformer having at least one primary winding and atleast one secondary winding; a switch assembly including a plurality ofswitching elements, the switch assembly being operatively connected tothe first input terminal, the second input terminal, the electricalelement, and the transformer primary winding, wherein the switchassembly is operative, in the event the first voltage is applied acrossthe first and second input terminals, to switchably apply the firstvoltage, the second voltage, and the reference voltage, across thetransformer primary winding to generate at least one third voltageacross the transformer secondary winding; a plurality of outputterminals; and a rectifier connected between the transformer secondarywinding and the output terminals.
 2. The DC-DC power converter of claim1 further including a filter connected between the rectifier and theoutput terminals.
 3. The DC-DC power converter of claim 1 wherein theelectrical element is selected from the group consisting of a capacitor,a battery, a solar cell, and a second DC-DC power converter.
 4. TheDC-DC power converter of claim 1 wherein the switch assembly comprises:first and second switching elements each having a first connection and asecond connection, wherein the first connection of the first switchingelement is connected to the first input terminal, the second connectionof the first switching element is connected to the first connection ofthe second switching element, and the second connection of the secondswitching element is connected to the second input terminal; third andfourth switching elements each having a first connection and a secondconnection, wherein the first connection of the third switching elementis connected to the first input terminal, the second connection of thethird switching element is connected to the first connection of thefourth switching element, and the second connection of the fourthswitching element is connected to the second input terminal; and whereina first connection of the transformer primary winding is connected tothe second connection of the first switching element, and a secondconnection of the transformer primary winding is connected to the secondconnection of the third switching element.
 5. The DC-DC power converterof claim 4 wherein the switch assembly further includes a fifthswitching element having a first connection and a second connection,wherein the first connection of the fifth switching element is connectedto a first connection of the at least one electrical element, and thesecond connection of the fifth switching element is connected to thefirst connection of the transformer primary winding.
 6. The DC-DC powerconverter of claim 1 wherein the at least one electrical elementincludes first and second electrical elements each having a firstconnection and a second connection, wherein the first connection of thefirst electrical element is connected to the first input terminal, thesecond connection of the first electrical element is connected to thefirst connection of the second electrical element, and the secondconnection of the second electrical element is connected to the secondinput terminal.
 7. The DC-DC power converter of claim 6 wherein thefirst and second electrical elements comprise respective capacitors. 8.The DC-DC power converter of claim 5 wherein the fifth switching elementincludes a plurality of switching sub-elements.
 9. The DC-DC powerconverter of claim 1 further including at least one inductor, theinductor and the transformer primary winding being serially connected,and wherein the switch assembly is operative, in the event the firstvoltage is applied across the first and second input terminals, toswitchably apply the first voltage, the second voltage, and thereference voltage across the inductor and the transformer primarywinding.
 10. The DC-DC power converter of claim 9 wherein at least oneof the plurality of switching elements is operative to provide the ZeroVoltage Switching (ZVS) property.
 11. The DC-DC power converter of claim1 wherein the transformer includes first and second primary windingseach having a first connection and a second connection, and the switchassembly comprises: first and second switching elements, the firstswitching element and the first transformer primary winding beingconnected across the first and second input terminals and forming afirst node, and the second switching element and the second transformerprimary winding being connected across the first and second inputterminals and forming a second node; and third and fourth switchingelements, the third switching element being connected across a firstconnection of the electrical element and the first node, and the fourthswitching element being connected across the first connection of theelectrical element and the second node.
 12. The DC-DC power converter ofclaim 11 further including a filter connected between the rectifier andthe output terminals.
 13. The DC-DC power converter of claim 11 whereineach of the third and fourth switching elements comprise a respectiveplurality of switching sub-elements.
 14. The DC-DC power converter ofclaim 11 wherein the transformer includes first and second secondarywindings.
 15. The DC-DC power converter of claim 1 further comprising afirst transformer and a second transformer each including a primarywinding and a secondary winding, each of the primary and secondarywindings having a first connection and a second connection, and theswitch assembly comprises: first and second switching elements, thefirst switching element and the primary winding of the first transformerbeing connected across the first and second input terminals and forminga first node, and the second switching element and the primary windingof the second transformer being connected across the first and secondinput terminals and forming a second node; and third and fourthswitching elements, the third switching element being connected across afirst connection of the electrical element and the first node, and thefourth switching element being connected across the first connection ofthe electrical element and the second node.
 16. The DC-DC powerconverter of claim 15 further including a filter connected between therectifier and the output terminals.
 17. A method of operating a DC-DCpower converter, comprising the steps of: in the event a first voltageis applied across first and second input terminals of the DC-DC powerconverter, providing a second voltage having a value between the firstvoltage value and a reference voltage value by at least one electricalelement connected to at least one of the first and second inputterminals; switchably applying the first voltage, the second voltage,and the reference voltage across a primary winding of a transformer by aswitch assembly, thereby generating at least one third voltage across asecondary winding of the transformer, the switch assembly beingoperatively connected to the first input terminal, the second inputterminal, the electrical element, and the transformer primary winding;and providing the at least one third voltage to a rectifier by thetransformer secondary winding, thereby generating a fourth rectifiedvoltage at output terminals of the DC-DC power converter.
 18. The methodof claim 17 further including the step of providing the fourth rectifiedvoltage to a filter by the rectifier, the filter being disposed betweenthe rectifier and the output terminals.
 19. The method of claim 17wherein the first providing step includes providing the second voltageby the at least one electrical element, the electrical element beingselected from the group consisting of a capacitor, a battery, and asecond DC-DC power converter.
 20. The method of claim 17 wherein theapplying step includes switchably applying at least the first voltageand the second voltage across the transformer primary winding by theswitch assembly, wherein the first voltage has a minimum value and amaximum value, the first voltage maximum value being less than or equalto two times the first voltage minimum value, and wherein the switchassembly includes a plurality of switching elements, the respectiveswitching elements operating according to at least one predeterminedduty ratio to allow the DC-DC power converter to operate with nodeadtime.
 21. The method of claim 17 wherein the applying step includesswitchably applying the first voltage, the second voltage, and thereference voltage across the transformer primary winding by the switchassembly, wherein the first voltage has a minimum value and a maximumvalue, the first voltage maximum value being greater than two times thefirst voltage minimum value, and wherein the switch assembly includes aplurality of switching elements, the respective switching elementsoperating according to at least one predetermined duty ratio to allowthe DC-DC power converter to operate with a predetermined amount ofdeadtime.
 22. The method of claim 17 wherein the applying step includesswitchably applying the first voltage, the second voltage, and thereference voltage across the transformer primary winding and at leastone inductor, the transformer primary winding and the at least oneinductor being serially connected.
 23. The method of claim 22 whereinthe applying step includes switchably applying the first voltage, thesecond voltage, and the reference voltage across the transformer primarywinding and the at least one inductor, the first voltage, the secondvoltage, and the reference voltage being switchably applied whilesatisfying the Zero Voltage Switching (ZVS) property in at least one ofa plurality of switching elements included in the switch assembly.
 24. ADC-DC power converter, comprising: a switch assembly including first andsecond input terminals, at least one electrical element connected to atleast one of the first and second input terminals, the at least oneelectrical element being operative, in the event a first voltage isapplied across the first and second input terminals, to provide a secondvoltage having a value between the first voltage value and a referencevoltage value, first and second output terminals, and a switchsubassembly including a plurality of switching elements, the switchsubassembly being operatively connected to the first input terminal, thesecond input terminal, and the electrical element, wherein the switchsubassembly is operative, in the event the first voltage is appliedacross the first and second input terminals, to switchably apply thefirst voltage, the second voltage, and the reference voltage across thefirst and second output terminals; and a second DC-DC power converteroperatively connected to the first and second output terminals of theswitch assembly, the second DC-DC power converter being configured toreceive the first voltage, the second voltage, and the reference voltageapplied across the first and second output terminals, and to generate atleast one third voltage.
 25. The DC-DC power converter of claim 24wherein the electrical element is selected from the group consisting ofa capacitor, a battery, a solar cell, and a third DC-DC power converter.26. The DC-DC power converter of claim 24 wherein the switch assemblycomprises: first and second switching elements each having a firstconnection and a second connection, the first connection of the firstswitching element being connected to the second connection of the secondswitching element to form a first node, and the first node beingconnected to a first connection of the electrical element; third andfourth switching elements, the third switching element being connectedbetween the first input terminal and the second connection of the firstswitching element, and the fourth switching element being connectedbetween the second input terminal and the first connection of the secondswitching element; and wherein the second connection of the firstswitching element and the first connection of the second switchingelement are connected to the first and second output terminals,respectively.
 27. The DC-DC power converter of claim 26 wherein thefirst and second switching elements comprise respective diodes.
 28. TheDC-DC power converter of claim 26 wherein the at least one electricalelement includes first and second electrical elements, the first andsecond electrical elements being serially connected to form the firstconnection of the at least one electrical element.
 29. The DC-DC powerconverter of claim 28 wherein the first and second electrical elementscomprise respective capacitors.
 30. The DC-DC power converter of claim24 wherein the second DC-DC power converter comprises an isolated DC-DCpower converter.
 31. The DC-DC power converter of claim 30 wherein theisolated DC-DC power converter comprises a dual-ended power converter.32. The DC-DC power converter of claim 31 wherein the dual-ended powerconverter is selected from the group consisting of a full-bridgeconverter, a half-bridge converter, and a push-pull converter.
 33. TheDC-DC power converter of claim 30 wherein the isolated DC-DC powerconverter comprises a dual-forward converter.
 34. The DC-DC powerconverter of claim 30 wherein the isolated DC-DC power converterincludes at least one primary winding, at least one secondary winding,and at least one inductor operatively coupled to the primary winding.35. The DC-DC power converter of claim 30 wherein the isolated DC-DCpower converter comprises a current-fed DC-DC power converter.
 36. TheDC-DC power converter of claim 35 wherein the current-fed DC-DC powerconverter is selected from the group consisting of a current-fedhalf-bridge DC-DC converter, a current-fed full-bridge DC-DC converter,a current fed push-pull DC-DC converter, and a current fed dual-forwardconverter.
 37. The DC-DC power converter of claim 30 wherein theisolated DC-DC power converter comprises a single-ended converter.
 38. Amethod of operating a DC-DC power converter, comprising the steps of: inthe event a first voltage is applied across first and second inputterminals of a switch assembly, providing a second voltage having avalue between the first voltage value and a reference value by at leastone electrical element included in the switch assembly; switchablyapplying the first voltage, the second voltage, and the referencevoltage across first and second output terminals of the switch assemblyby a switch subassembly, the switch subassembly including a plurality ofswitching elements and being operatively connected to the first inputterminal, the second input terminal, and the electrical element;receiving the first voltage, the second voltage, and the referencevoltage applied across the first and second output terminals of theswitch assembly by a second DC-DC power converter; and generating atleast one third voltage by the second DC-DC power converter.
 39. Themethod of claim 38 wherein the providing step includes providing thesecond voltage by the at least one electrical element, the electricalelement being selected from the group consisting of a capacitor, abattery, and a third DC-DC power converter.
 40. The method of claim 38wherein the receiving and generating steps include receiving the firstvoltage, the second voltage, and the reference voltage, and generatingthe at least one third voltage, by the second DC-DC power converter, thesecond DC-DC power converter comprising an isolated DC-DC powerconverter.
 41. The method of claim 40 wherein the receiving andgenerating steps include receiving the first voltage, the secondvoltage, and the reference voltage, and generating the at least onethird voltage, by the isolated DC-DC power converter, the isolated DC-DCpower converter comprising a dual-ended power converter.
 42. The methodof claim 38 wherein the applying step includes switchably applying atleast the first voltage and the second voltage across the first andsecond output terminals of the switch assembly, wherein the firstvoltage has a minimum value and a maximum value, the first voltagemaximum value being less than or equal to two times the first voltageminimum value, and wherein the respective switching elements operateaccording to at least one predetermined duty ratio to allow the DC-DCpower converter to operate with no deadtime.
 43. The method of claim 38wherein the applying step includes switchably applying the firstvoltage, the second voltage, and the reference voltage across the firstand second output terminals of the switch assembly, wherein the firstvoltage has a minimum value and a maximum value, the first voltagemaximum value being greater than two times the first voltage minimumvalue, and wherein the respective switching elements operate accordingto at least one predetermined duty ratio to allow the DC-DC powerconverter to operate with a predetermined amount of deadtime.
 44. ADC-DC power converter, comprising: a switch assembly including first andsecond input terminals, at least one electrical element connected to atleast one of the first and second input terminals, the at least oneelectrical element being operative, in the event a first voltage isapplied across the first and second input terminals, to provide a secondvoltage having a value between the first voltage value and a referencevoltage value, first and second output terminals, and a switchsubassembly including a plurality of switching elements, the switchsubassembly being operatively connected to the first input terminal, thesecond input terminal, and the electrical element, wherein the switchsubassembly is operative, in the event the first voltage is appliedacross the first and second input terminals, to switchably apply thefirst voltage, the second voltage, and the reference voltage across thefirst and second output terminals; and a filter operatively connected tothe first and second output terminals of the switch assembly, the filterbeing configured to receive the switchably applied first, second, andreference voltages, and to generate at least one third filtered voltage.45. The DC-DC power converter of claim 44 further including a secondDC-DC power converter operatively connected to the filter, the secondDC-DC power converter being configured to receive the at least one thirdfiltered voltage, and to generate at least one fourth voltage.
 46. TheDC-DC power converter of claim 44 wherein the electrical element isselected from the group consisting of a capacitor, a battery, and athird DC-DC power converter.
 47. The DC-DC power converter of claim 44wherein the switch assembly comprises: first and second switchingelements each having a first connection and a second connection, thefirst connection of the first switching element being connected to thesecond connection of the second switching element to form a first node,and the first node being connected to a first connection of theelectrical element; third and fourth switching elements, the thirdswitching element being connected between the first input terminal andthe second connection of the first switching element, and the fourthswitching element being connected between the second input terminal andthe first connection of the second switching element; and wherein thesecond connection of the first switching element and the firstconnection of the second switching element are connected to the firstand second output terminals, respectively.
 48. The DC-DC power converterof claim 47 wherein the first and second switching elements compriserespective diodes.
 49. The DC-DC power converter of claim 47 wherein theat least one electrical element includes first and second electricalelements, the first and second electrical elements being seriallyconnected to form the first connection of the at least one electricalelement.
 50. The DC-DC power converter of claim 49 wherein the first andsecond electrical elements comprise respective capacitors.
 51. The DC-DCpower converter of claim 44 wherein the second DC-DC power convertercomprises an isolated DC-DC power converter.
 52. The DC-DC powerconverter of claim 51 wherein the isolated DC-DC power convertercomprises a dual-ended power converter.
 53. The DC-DC power converter ofclaim 52 wherein the dual-ended power converter is selected from thegroup consisting of a full-bridge converter, a half-bridge converter, adual-forward converter, and a push-pull converter.
 54. The DC-DC powerconverter of claim 51 wherein the isolated DC-DC power convertercomprises a dual-forward converter.
 55. The DC-DC power converter ofclaim 51 wherein the isolated DC-DC power converter includes at leastone primary winding, at least one secondary winding, and at least oneinductor operatively coupled to the primary winding.
 56. The DC-DC powerconverter of claim 51 wherein the isolated DC-DC power convertercomprises a current-fed DC-DC power converter.
 57. The DC-DC powerconverter of claim 56 wherein the current-fed DC-DC power converter isselected from the group consisting of a current-fed half-bridge DC-DCconverter, a current-fed full-bridge DC-DC converter, a current fedpush-pull DC-DC converter, and a current fed dual-forward converter. 58.The DC-DC power converter of claim 51 wherein the isolated DC-DC powerconverter comprises a single-ended converter.
 59. A method of operatinga DC-DC power converter, comprising the steps of: in the event a firstvoltage is applied across first and second input terminals of a switchassembly, providing a second voltage having a value between the firstvoltage value and a reference value by at least one electrical elementincluded in the switch assembly; switchably applying the first voltage,the second voltage, and the reference voltage across first and secondoutput terminals of the switch assembly by a switch subassembly, theswitch subassembly including a plurality of switching elements and beingoperatively connected to the first input terminal, the second inputterminal, and the electrical element; receiving the first voltage, thesecond voltage, and the reference voltage applied across the first andsecond output terminals of the switch assembly by a filter; andgenerating at least one third filtered voltage by the filter.
 60. Themethod of claim 59 further including the steps of receiving the at leastone third filtered voltage by a second DC-DC power converter; andgenerating at least one fourth voltage by the second DC-DC powerconverter.
 61. The method of claim 59 wherein the providing stepincludes providing the second voltage by the at least one electricalelement, the electrical element being selected from the group consistingof a capacitor, a battery, and a third DC-DC power converter.
 62. Themethod of claim 59 wherein the second receiving step includes receivingthe at least one third filtered voltage by the second DC-DC powerconverter, the second DC-DC power converter comprising an isolated DC-DCpower converter.
 63. The method of claim 62 wherein the second receivingstep includes receiving the at least one third filtered voltage by theisolated DC-DC power converter, the isolated DC-DC power convertercomprising a dual-ended power converter.
 64. The method of claim 62wherein the second receiving step includes receiving the at least onethird filtered voltage by the isolated DC-DC power converter, theisolated DC-DC power converter comprising a dual forward converter. 65.The method of claim 59 wherein the applying step includes switchablyapplying at least the first voltage and the second voltage across thefirst and second output terminals of the switch assembly, wherein thefirst voltage has a minimum value and a maximum value, the first voltagemaximum value being less than or equal to two times the first voltageminimum value, and wherein the respective switching elements operateaccording to at least one predetermined duty ratio to allow the DC-DCpower converter to operate with no deadtime.
 66. The method of claim 60wherein the applying step includes switchably applying at least thefirst voltage and the second voltage across the first and second outputterminals of the switch assembly, and wherein the respective switchingelements operate according to at least one predetermined duty ratio toallow the DC-DC power converter to operate with no deadtime.
 67. Themethod of claim 59 wherein the applying step includes switchablyapplying the first voltage, the second voltage, and the referencevoltage across the first and second output terminals of the switchassembly, wherein the first voltage has a minimum value and a maximumvalue, the first voltage maximum value being greater than two times thefirst voltage minimum value, and wherein the respective switchingelements operate according to at least one predetermined duty ratio toallow the DC-DC power converter to operate with a predetermined amountof deadtime.